The Clarinet Tonehole as a Two-Port Junction

The clarinet tonehole model developed by Keefe [242] is
parametrized in terms of series and shunt resistance and reactance, as
shown in Fig. 9.43. The *transmission
matrix* description of this two-port is given by the product of the
transmission matrices for the series impedance
, shunt
impedance
, and series impedance
, respectively:

where all quantities are written in the frequency domain, and the impedance parameters are given by

(open-hole shunt impedance) | |||

(closed-hole shunt impedance) | (10.51) | ||

(open-hole series impedance) | |||

(closed-hole series impedance) |

where is the wave impedance of the tonehole entrance,

where is the radius of the main bore. The closed-tonehole height can be estimated as [242]

where is the physical tonehole chimney height at its center.

Note that the specific resistance of the open tonehole, , is the only real impedance and therefore the only source of wave energy loss at the tonehole. It is given by [242]

ln

where is the radius of curvature of the tonehole, is the viscous boundary layer thickness which expressible in terms of the shear viscosity of air as

and is the real part of the propagation wavenumber (or minus the imaginary part of complex spatial frequency ). In [241], for the large-tube limit (

where is the adiabatic gas constant for air [321], is the thermal conductivity of air, and is the specific heat of air at constant pressure. In [241], the following values are given for air at Kelvin ( C), and valid within degrees of that temperature:

where

can be interpreted as times the ratio of the tonehole radius to the viscous boundary layer thickness [241]. The constant is referred to as the Prandtl number, and is the shear viscosity coefficient [241]. In [71], it is noted that is greater than under practical conditions in musical acoustics, and so it is therefore sufficient to keep only the first and second-order terms in the expression above for .

The open-hole effective length , assuming no pad above the hole, is given in [242] as

See [242] for the case in which a pad lies above the open hole. In [408], a unified tonehole model is given which supports continuous opening and closing of the tonehole.

For implementation in a digital waveguide model, the lumped parameters above must be converted to scattering parameters. Such formulations of toneholes have appeared in the literature: Vesa Välimäki [511,504] developed tonehole models based on a ``three-port'' digital waveguide junction loaded by an inertance, as described in Fletcher and Rossing [144], and also extended his results to the case of interpolated digital waveguides. It should be noted in this context, however, that in the terminology of Appendix C, Välimäki's tonehole representation is a loaded 2-port junction rather than a three-port junction. (A load can be considered formally equivalent to a ``waveguide'' having wave impedance given by the load impedance.) Scavone and Smith [405] developed digital waveguide tonehole models based on the more rigorous ``symmetric T'' acoustic model of Keefe [242], using general purpose digital filter design techniques to obtain rational approximations to the ideal tonehole frequency response. A detailed treatment appears in Scavone's CCRMA Ph.D. thesis [409]. This section, adapted from [467], considers an exact translation of the Keefe tonehole model, obtaining two one-filter implementations: the ``shared reflectance'' and ``shared transmittance'' forms. These forms are shown to be stable without introducing an approximation which neglects the series inertance terms in the tonehole model.

By substituting
in (9.53) to convert spatial
frequency to temporal frequency, and by substituting

(10.52) | |||

(10.53) |

for , into (9.51) to convert physical variables to wave variables, ( is the bore wave impedance), we may solve for the outgoing waves in terms of the incoming waves . Mathematica code for obtaining the general conversion formula from lumped parameters to scattering parameters is as follows:

Clear["t*", "p*", "u*", "r*"] transmissionMatrix = {{t11, t12}, {t21, t22}}; leftPort = {{p2p+p2m}, {(p2p-p2m)/r2}}; rightPort = {{p1p+p1m}, {(p1p-p1m)/r1}}; Format[t11, TeXForm] := "{T_{11}}" Format[p1p, TeXForm] := "{P_1^+}" ... (etc. for all variables) ... TeXForm[Simplify[Solve[leftPort == transmissionMatrix . rightPort, {p1m, p2p}]]]The above code produces the following formulas:

Substituting relevant values for Keefe's tonehole model, we obtain, in matrix notation,

We thus obtain the scattering formulation depicted in Fig. 9.44, where

is the

is the

The approximate forms in (9.57) and (9.58) are obtained by neglecting the negative series inertance which serves to adjust the effective length of the bore, and which therefore can be implemented elsewhere in the interpolated delay-line calculation as discussed further below. The open and closed tonehole cases are obtained by substituting and , respectively, from (9.53).

In a manner analogous to converting the four-multiply Kelly-Lochbaum (KL) scattering junction [247] into a one-multiply form (cf. (C.60) and (C.62) on page ), we may pursue a ``one-filter'' form of the waveguide tonehole model. However, the series inertance gives some initial trouble, since

instead of zero as in the KL junction. In the scattering formulas (C.100) and (C.101) on page for the general loaded waveguide junction, the reflectance seen on any branch is always the transmittance from that branch to any other branch minus .

into the basic scattering relations (9.56), and factoring out , we obtain, in the frequency domain,

(10.58) |

and, similarly,

(10.59) |

The resulting tonehole implementation is shown in Fig. 9.45. We call this the ``shared reflectance'' form of the tonehole junction.

In the same way, an alternate form is obtained from the substitution

which yields the ``shared transmittance'' form:

(10.60) | |||

(10.61) |

shown in Fig. 9.46.

Since , it can be neglected to first order, and , reducing both of the above forms to an approximate ``one-filter'' tonehole implementation.

Since is a pure negative reactance, we have

In this form, it is clear that is a first-order

We now see precisely how the negative series inertance
provides a
*negative, frequency-dependent, length correction* for the bore. From
(9.63),
the phase delay of
can be computed as

Thus, the negative delay correction goes to zero with frequency , series tonehole length , tonehole impedance , or main bore admittance .

In practice, it is common to combine all delay corrections into a single ``tuning allpass filter'' for the whole bore [432,208]. Whenever the desired allpass delay goes negative, we simply add a sample of delay to the desired allpass phase-delay and subtract it from the nearest delay. In other words, negative delays have to be ``pulled out'' of the allpass and used to shorten an adjacent interpolated delay line. Such delay lines are normally available in practical modeling situations.

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